Voltage regulator with load compensation

ABSTRACT

A voltage regulation system provides a relatively stable voltage source without introducing the typical costs of a ground buffer. The disclosed voltage regulation system includes a voltage regulator that is operative to detect a change of the load current and regulate a current bypass mechanism to stabilize a total supply current. For example, the voltage regulator includes a current sensor and a current compensation circuit. The current sensor is configure to generate a current compensation signal based on the load current change, whereas the current compensation circuit is configured to adjust a bypass current in response to the current compensation signal. As a result, the bypass current dynamically compensates the load current change such that the ground voltage of a variable load becomes relatively stable over a range of load currents.

CROSS REFERENCE TO RELATED APPLICATIONS

Under 35 U.S.C. §§119(e), 120, this continuation application claimspriority to and benefits of U.S. patent application Ser. No. 14/848,837,filed on Sep. 9, 2015, which claims the benefits of and priority to U.S.Provisional Application No. 62/048,482 filed on Sep. 10, 2014. Theentirety of the above referenced applications is hereby incorporatedherein by reference.

BACKGROUND

Voltage regulators are used for providing voltage sources for operatinganalog circuitries. These voltage sources may include supply voltages(e.g., VDD) and reference voltages (e.g., VREF). A load circuit thatreceives the voltage source may experience a change of impedance duringoperations, such that the corresponding load current may be operationdependent. For instance, a digital-to-analog conversion (DAC) circuittypically includes a resistor network that has a variable resistancedepending on a digital code. The DAC circuit operates to convert thedigital code to an analog signal by manipulating the resistor networkunder a reference voltage. When the variable resistance changesaccording to the digital code, the resistor network may drain adifferent amount of load current to a ground source. In the likelyscenario where the ground source includes parasitic elements, thechanging load current will cause the ground voltage to fluctuate. Thefluctuating ground voltage directly impacts the stability of thereference voltage. As a result, the performance and reliability of theDAC circuit may become code-dependent.

To alleviate the code-dependency of the DAC circuit, attempts have beenmade in the past to use a ground buffer in conjunction with a voltageregulator. The ground buffer operates to stabilize the ground voltage ofthe resistor network by means of a feedback control mechanism. However,the ground buffer generally increases the design complexity, the powerconsumption, and the size of the overall circuit. The deployment ofground buffers thus becomes infeasible in systems with stringent designconstraints. Accordingly, there is a need for a voltage regulator thatcan provide a relatively stable voltage source without incurring thedesign costs of a ground buffer.

SUMMARY

The present disclosure describes a voltage regulation system thatprovides a relatively stable voltage source without introducing thetypical costs of a ground buffer. The disclosed voltage regulationsystem includes a voltage regulator that is operative to detect a changeof the load current and regulate a current bypass mechanism in responseto the detected load current change. The current bypass mechanismdynamically compensates the load current change, thereby stabilizing atotal ground current. As a result, the ground voltage of a variable loadbecomes relatively stable over a range of load currents.

In one implementation, for example, the present disclosure describes areference voltage regulation (RVR) circuit for supplying a referencevoltage across a variable load. The RVR circuit includes a first outputnode, a second output node, a current compensation circuit, and acurrent sensor. The first output node is configured to deliver an outputcurrent to the variable load. The second output node is configured tocollect the output current from the variable load. The currentcompensation circuit is coupled between the first and second outputnodes. The current compensator circuit has a control terminal that isconfigured to receive a current compensation signal for adjusting acompensation current across the first and second output nodes via thecurrent compensator circuit. The current sensor is coupled with thefirst and second output nodes to sense a change of the output current.As a result of the sensing, the current sensor is configured to generatethe current compensation signal based on the sensed change of the outputcurrent.

In another implementation, for example, the present disclosure describesa reference voltage regulation (RVR) circuit for supplying a referencevoltage across a variable load. The RVR circuit includes a first outputnode, a second output node, a current compensation circuit, and acurrent sensor with a supply current path and a monitoring current path.The first output node configured to deliver an output current to thevariable load. The second output node configured to collect the outputcurrent from the variable load. The current compensation circuit iscoupled between the first and second output nodes. The currentcompensator circuit has a control terminal that is configured to receivea current compensation signal for adjusting a compensation currentacross the first and second output nodes via the current compensatorcircuit. The supply current path has a supply output node that iscoupled with the first output node to deliver a supply currentsustaining the output current and the compensation current. Themonitoring current path is configured to deliver a monitoring currentresponsive to the sensed change of the output current. Moreover, themonitoring current path has a monitoring output node to deliver thecurrent compensation signal based on the monitoring current.

In another implementation, for example, the present disclosure describesa reference voltage regulation (RVR) circuit for supplying a referencevoltage across a variable load. The RVR circuit includes a first outputnode, a second output node, a current compensation circuit, and acurrent sensor with a first sensing stage and a second sensing stage.The first output node configured to deliver an output current to thevariable load. The second output node configured to collect the outputcurrent from the variable load. The current compensation circuit iscoupled between the first and second output nodes. The currentcompensator circuit has a control terminal that is configured to receivea current compensation signal for adjusting a compensation currentacross the first and second output nodes via the current compensatorcircuit. The first sensing stage is coupled with the first output nodeto sense the change of the output current. The first sensing stage isconfigured to generate a current sense signal responding positively tothe sensed change of the output current. The second sensing stage iscoupled with the first sensing stage to receive the current sensesignal. The second sensing stage is configured to generate the currentcompensation signal based on a comparison between the current sensesignal and a predetermined set voltage.

In yet another implementation, for example, the present disclosuredescribes a digital-to-analog conversion (DAC) system that includes avariable resistance network, a current compensation circuit, and acurrent sensor. The variable resistance network has a high referencenode and a low reference node. The variable resistance network isconfigured to conduct a load current from the high reference node to thelow reference node such that the load current is adjustable based on adigital code. The current compensation circuit is coupled between thehigh and low reference nodes. The current compensator circuit has acontrol terminal that is configured to receive a current compensationsignal for adjusting a compensation current bypassing the variableresistance network. The current sensor is coupled with the high and lowreference nodes to sense a change of the load current. As a result ofthe sensing, the current sensor is configured to generate the currentcompensation signal based on the sensed change of the load current.

DRAWING DESCRIPTIONS

FIG. 1 shows a schematic view of an exemplary voltage regulation systemaccording to an aspect of the present disclosure.

FIG. 2 shows a schematic view of an exemplary voltage regulation circuitaccording to an aspect of the present disclosure.

FIG. 3 shows a schematic view of an exemplary voltage regulation circuitaccording to another aspect of the present disclosure.

FIG. 4 shows a schematic view of an exemplary voltage regulation systemwith two sensing stages according to an aspect of the presentdisclosure.

FIG. 5 shows a schematic view of an exemplary digital-to-analogconversion (DAC) circuit according to an aspect of the presentdisclosure.

Like reference symbols in the various drawings indicate like elements.Details of one or more implementations of the present disclosure are setforth in the accompanying drawings and the description below. Thefigures are not drawn to scale and they are provided merely toillustrate the disclosure. Specific details, relationships, and methodsare set forth to provide an understanding of the disclosure. Otherfeatures and advantages may be apparent from the description anddrawings, and from the claims.

DETAILED DESCRIPTION

FIG. 1 shows a schematic view of an exemplary voltage regulation system100 according to an aspect of the present disclosure. The voltageregulation system 100 includes a voltage regulation circuit 120 and avariable load circuit 110. In operation, the voltage regulation circuit120 provides a relatively stable source of voltage across the variableload circuit 110. The source of voltage may include a supply voltage(e.g., VDD) and/or a reference voltage (e.g., V_(REF)). The voltageregulation circuit 120 includes a current bypass control mechanism todynamically adjust a compensation current 133 that bypasses the variableload circuit 110. When the impedance of the variable load circuit 110changes, the load current 113 fluctuates. The compensation current 133generally compensates the fluctuations in the load current 113, therebystabilizing a ground current 114. The stabilization of the groundcurrent 114 also helps stabilize the voltage across the variable loadcircuit 110.

The voltage regulation circuit 120 includes an input node 124, a firstoutput node 121, and a second output node 123. In the event that thevoltage regulation circuit 120 regulates a supply voltage, the inputnode 124 is configured to receive the supply voltage for regulation.Alternatively, in the event that the voltage regulation circuit 120regulates a reference voltage, the input node 124 is configured toreceive the reference voltage (V_(REF)) for regulation. The voltageregulation circuit 120 monitors and adjusts its output at the firstoutput node 121 and the second output node 123 so as to maintainthereacross a potential difference that corresponds to the voltagereceived at the input node 124. For instance, the input node 124 mayreceive a reference voltage (V_(REF)) relative to a ground voltage. Theinput reference voltage (V_(REF)) may be generated by a referencevoltage generator (e.g., a voltage divider circuit).

The voltage regulation circuit 120 monitors the load current 113 of thevariable load circuit 110 and adjusts the compensation current 133accordingly to maintain a relatively constant potential differenceacross the first output node 121 and the second output node 123. Assuch, the first output node 121 is regulated at a high reference voltage(V_(REFH)) whereas the second output node 123 is regulated at a lowreference voltage (V_(REFL)). The difference between the high referencevoltage (V_(REFH)) and the low reference voltage (V_(REFL)) correspondsto the input reference voltage (V_(REF)).

More specifically, the first output node 121 is coupled to a highreference node 111 of the variable load circuit 110, whereas the secondoutput node 123 is coupled to a low reference node 112 of the variableload circuit 110. In this configuration, the first output node 121delivers an output current 122 to the variable load circuit 110, whereasthe second output node 123 collects the delivered output current (i.e.,the load current) 113 from the variable load circuit 110. The variableload circuit 110 may have output impedance that varies according to oneor more operations of the variable load circuit 110. As the voltageregulation circuit 120 maintains a relatively stable potential acrossthe high reference node 111 and the low reference node 112, the changingoutput impedance may introduce a change in output current 122 across thevariable load circuit 110. In one implementation, for instance, thevariable load circuit 110 may be a digital-to-analog conversion (DAC)circuit which includes a variable resistance network that is configuredto conduct a load current 113 based on its load impedance. Because theload impedance changes according to a digital input code, the loadcurrent 113 is dependent on the digital input code.

To serve as a voltage regulation means, the voltage regulation circuit120 includes a current compensation circuit 130 and a current sensor140. The current compensation circuit 130 serves as a bypassing meansfor dynamically adjusting a compensation current 133 that bypasses thevariable load circuit 110. The current sensor 140 serves as a feedbackmeans for tracking the load current 113 and controlling the currentcompensation circuit 130 to compensate changes in the load current 113.Working in conjunction with each other, the current compensation circuit130 and the current sensor 140 sustain a supply current 174 and a groundcurrent 114 at a relatively constant level.

The current compensation circuit 130 is coupled between the first outputnode 121 and the second output node 123. The current compensationcircuit 130 includes a control terminal 131 that is configured toreceive a current compensation signal 132 from the current sensor 140.The current compensation signal 132 directs the current compensationcircuit 130 to adjust the compensation current 133. The compensationcurrent 133 is conducted across the first output node 121 and the secondoutput node 123 to bypass the variable load circuit 110. By adjustingthe compensation current 133, the current compensation circuit 130 helpssustain a relatively constant ground current 114 over a range of loadcurrent 113. In return, the relatively constant ground current 114maintains a relatively constant ground voltage above the groundparasitic resistor 115. The relatively constant ground voltagestabilizes the potential difference across the high reference node 111and the low reference node 112 of the variable load circuit 110.

The current sensor 140 is coupled with the first output node 121 and thesecond output node 122 to sense a change of the output current 122,which is induced by a change of the load current 113 due to a change ofthe load impedance of the variable load circuit 110. The current sensor140 is configured to generate the current compensation signal 132 basedon the sensed change of the output current 122. In general, the currentsensor 140 is configured to adjust the current compensation signal 132to stabilize a supply current 174, which is distributed as thecompensation current 133 and the output current 122 at the first outputnode 121. The compensation current 133 and the output current 122 rejoinat the second output node 123 to form the ground current 114. Bystabilizing the supply current 174, the current sensor 140 alsostabilizes the ground current 114 and the ground voltage across theground parasitic resistor 115. For example, upon sensing a reduction ofthe output current 122, the current sensor 140 is configured to adjustthe current compensation signal 132 to increase the compensation current133. Alternatively, upon sensing an increment of the output current 122,the current sensor 140 is configured to adjust the current compensationsignal 132 to reduce the compensation current 133.

In one configuration, the current sensor 140 includes a VREF comparisoncircuit 150, a monitoring current path 160, and a supply current path170. The VREF comparison circuit 150 serves as a tracking means fortracking a change of voltage at the first output node 121. This changeof voltage indicates a transient change of output current 122 due to achange of output impedance of the variable load circuit 110. The VREFcomparison circuit 150 includes a first input 152, a second input 154, afirst output 156, and a second output 158. The first input 152 iscoupled with the first output node 121 via a feedback path 125. To thatend, the first input 152 may share a DC voltage with the first outputnode 121. The second input 154 is coupled to the input node 124 of thevoltage regulation circuit 120 for receiving the input reference voltage(V_(REF)).

In general, the VREF comparison circuit 150 is configured to track thehigh reference voltage (V_(REFH)) at the first output node 121 bycomparing the high reference voltage (V_(REFH)) to the input referencevoltage (V_(REF)) receives by the input node 124 of the voltageregulation circuit 120. Based on this comparison, the VREF comparisoncircuit 150 generates a first feedback control signal 141 and a secondfeedback ground signal 142. The first feedback signal 141 is output bythe first output 156, and it is then fed to the monitoring current path160 and the supply current path 170. The second feedback signal 142 isoptionally delivered at the second output 158, and it may be fed to thesupply current path 170 in conjunction with the first feedback controlsignal 141. When the VREF comparison circuit 150 operates within itsoperation range, the first and second feedback signals 141 and 142 mayhave a substantially linear relationship with a difference between theinput reference voltage (V_(REF)) and the high reference voltage(V_(REFH)).

The monitoring current path 160 is coupled with the VREF comparisoncircuit 150 to receive the first feedback control signal 141. In oneimplementation, for instance, the monitoring current path 160 includes amonitoring input node 161 to receive the first feedback control signal141 from the first output 156 of the VREF comparison circuit 150. Inresponse to the first feedback control signal 141, the monitoringcurrent path 160 generates and delivers a monitoring current 165.Because the first feedback control signal 141 represents a sensed changeof the output current 122, the monitoring current 165 is responsive tothe sensed change of the output current 122 as well. The monitoringcurrent 165 drives the current compensation signal 132 such that thecurrent compensation signal 132 is also responsive to the sensed changeof output current 122. The monitoring current path 160 includes amonitoring output node 162 for delivering the current compensationsignal 132.

The supply current path 170 is coupled with the VREF comparison circuit150 to receive the first feedback control signal 141 and optionally thesecond feedback control signal 142. In one implementation, for instance,the supply current path 170 includes a first supply input node 171coupled with the first output 156 to receive the first feedback controlsignal 141, and a second supply input node 172 coupled with the secondoutput 158 to receive the second feedback control signal 142. Inresponse to the first feedback control signal 141 and optionally to thesecond feedback control signal 142, the supply current path 170 deliversthe supply current 174 to sustain the output current 122 and thecompensation current 133. Although the supply current path 170 maintainsa relatively stable supply current 174 for DC operations, the supplycurrent path 170 may also be adjusted by the first and second feedbackcontrol signals 141 and 142 to provide a transient response to a changeof output current 122. The transient response in the supply current 174serves to maintain the high reference voltage (V_(REFH)) at the firstoutput node 121 while the current compensation circuit 130 is respondingto the changing output current 122.

FIG. 2 shows a schematic view of an exemplary voltage regulation circuit120 according to an aspect of the present disclosure. The voltageregulation circuit 120 as shown in FIG. 2 provides a specificimplementation of the voltage regulation circuit 120 as shown in FIG. 1.While the circuitry in FIG. 2 further elaborates the structure andoperations of the voltage regulation circuit 120 in FIG. 1, it does notlimit or restrict the description of FIG. 1.

The VREF comparison circuit 150 includes an amplifier 251 and an outputstage 255. The amplifier 251 has a negative input 152 and a positiveinput 154. The negative input 152 is coupled with the first output node121 to receive the high reference voltage (V_(REFH)) via the feedbackpath 125, whereas the positive input 154 is coupled with input node 124to receive the input reference voltage (V_(REF)). In an exemplaryconfiguration, the amplifier 251 detects a difference between the inputreference voltage (V_(REF)) and the high reference voltage (V_(REFH))and amplifies the detected difference under a substantially linearfunction, for instance: Vo=Ao(V_(plus)−V_(minus)), where Vo is theamplification output at the output node 254 and Ao is the gain of thelinear function. The amplifier 251 may include one or more differentialamplifier in one implementation. In another implementation, theamplifier 251 may include one or more operational amplifier. In yetanother implementation, the amplifier 251 can include other types ofamplification circuits that are suitable for performing the functions asdescribed above.

Depending on the configuration of the monitoring current path 160 andthe supply current path 170, the VREF comparison circuit 150 may includean output stage 255. In the event that the amplifier 251 is anoperational amplifier, the output stage 255 may be an operational outputstage with a class AB bias. The output stage 255 is configured toprocess the amplification output received from the output node 254 andgenerate the first and second feedback control signals 141 and 142. Ingenerate, first and second feedback control signals 141 and 142 areadjusted proportionally to the change of the output current 122 and forthe function of maintaining a relatively stable high reference voltage(V_(REFH)) at the first output node 121.

The supply current path 170 includes a p-channel transistor 271 and ann-channel transistor 276. The p-channel transistor 271 includes a sourcenode 272, a gate node 273, and a drain node 274. The source node 272 iscoupled to an internal voltage supply source 201, such as VDD or VCC.The gate node 273 is coupled to the first supply input node 171 toreceive the first feedback control signal 141. Based on a differencebetween the internal voltage supply source 201 and the voltage of thefirst feedback control signal 141, the p-channel transistor 271establishes a gate-to-source voltage V_(GS), which in turns controls theamount of supply current 174 the p-channel transistor 271 conducts.Because the first feedback control signal 141 is responsive to thechange of output current 122, the adjustment of the supply current 174is also responsive to the change of the output current 122. Thep-channel transistor 271 conducts the supply current 174 from its sourcenode 272 to its drain node 274, which is coupled to the supply outputnode 173 for delivering the supply current 174. Thus, the p-channeltransistor 271 is a part of a supply current feedback loop, within whichthe supply current 174 is monitored and regulated by the VREF comparisoncircuit 150 and the supply current path 170. The p-channel transistor271 may include one or more PMOS transistors in one implementation.Alternatively, the p-channel transistor 271 may include other types oftransistors that are suitable for performing the functions as describedabove.

The n-channel transistor 276 serves as a bias component to maintain acertain voltage across the p-channel transistor 271 such that thep-channel transistor 271 may have a proper amount of drain-to-sourcevoltage V_(DS) _(_) ₁ to drive the supply current 174. By conducting asource current (or quiescent current), the n-channel transistor 276establishes its own drain-to-source voltage V_(DS) _(_) ₂ across itsdrain node 277 and its source node 279. This drain-to-source voltageV_(DS) _(_) ₂ contributes to the stability of the high reference voltage(V_(REFH)) as the drain node 277 is coupled to the first output node121. The n-channel transistor 276 includes a gate node 278, which iscoupled to the second supply input node 172 to receive the secondfeedback control signal 142. Using the second feedback control signal142, the VREF comparison circuit 150 controls the amount of sourcecurrent conducted by the n-channel transistor 276. By adjusting the gatevoltage at the gate node 278, the VREF comparison circuit 150 alsoregulates the drain voltage at the drain node 277, which in turns helpsmaintain a relatively constant high reference voltage (V_(REFH)). Thus,the n-channel transistor 276 is a part of a high reference voltage(V_(REFH)) feedback loop, within which the high reference voltage(V_(REFH)) is monitored and regulated by the VREF comparison circuit 150and the supply current path 170. The n-channel transistor 276 mayinclude one or more NMOS transistors in one implementation.Alternatively, the n-channel transistor 276 may include other types oftransistors that are suitable for performing the functions as describedabove.

The monitoring current path 160 includes a p-channel transistor 261, aset resistor 266, and a set voltage (VSET) comparison circuit 267. Thep-channel transistor 261 includes a source node 262, a gate node 263,and a drain node 264. The source node 262 is coupled to the internalvoltage supply source 201. The gate node 263 is coupled to the firstsupply input node 161 to receive the first feedback control signal 141.Based on a difference between the internal voltage supply source 201 andthe voltage of the first feedback control signal 141, the p-channeltransistor 261 establishes a gate-to-source voltage (V_(GS)), which inturns controls the amount of monitoring current 165 the p-channeltransistor 261 conducts.

Because the first feedback control signal 141 is responsive to thechange of output current 122, the adjustment of the monitoring current165 is also responsive to the change of the output current 122.Moreover, the p-channel transistor 261 and the p-channel transistor 271are arranged in a mirror configuration such that both transistors arebiased with substantially the same amount of gate-to-source voltage(V_(GS)). In a configuration that the p-channel transistor 261 and thep-channel transistor 271 have similar electron mobility, the monitoringcurrent 165 is proportional to, and thus keeps track of, the supplycurrent 174. Thus, the monitoring current 165 is sensitive andresponsive to the change of the supply current 174, which in part can beattributed by a change of the output current 122. The p-channeltransistor 261 may include one or more PMOS transistors in oneimplementation. Alternatively, the p-channel transistor 261 may includeother types of transistors that are suitable for performing thefunctions as described above

The monitoring current 165 is generally less than the supply current 174because the monitoring current 165 is not used for supplying the loadcurrent 113. Thus, the size of the p-channel transistor 261 is typicallysmaller than that of the p-channel transistor 271. The p-channeltransistor 261 conducts the monitoring current 165 from its source node262 to its drain node 264, which is coupled to a set node 265. The setresistor 266 is coupled to the set node 265 to receive the monitoringcurrent 165. By conducting the monitoring current 165, the set resistor266 establishes a monitoring set voltage (V_(SETM)) at the set node 165.Because the monitoring set voltage (V_(SETM)) is a function of themonitoring current 165, the monitoring set voltage (V_(SETM)) issensitive and responsive to the supply current 174, which in part can beattributed by a change of the output current 122.

The VSET comparison circuit 267 is configured to compare the monitoringset voltage (V_(SETM)) with a predetermined set voltage (V_(SETP)) so asto generate the current compensation signal 132. The VSET comparisoncircuit 267 serves as a tracking means for tracking a change in themonitoring current 165, which is indicative of a change in the supplycurrent 174 as well as the output current 122. The VSET comparisoncircuit 267 includes a first input 268 (which can be a negative input),a second input 269 (which can be a positive input), and a comparisonoutput 260. The first input 268 is coupled to the set node 265 toreceive the monitoring set voltage (V_(SETM)), whereas the second input269 is coupled to a voltage source to receive the predetermined setvoltage (V_(SETP)). The VSET comparison circuit 267 delivers the currentcompensation signal 132 to the monitoring output node 162 via thecomparison output 260. When the VSET comparison circuit 267 operateswithin its operation range, the current compensation signal 132 may havea substantially linear relationship with a difference between thepredetermined set voltage (V_(SETP)) and the monitoring set voltage(V_(SETM)).

Like the VREF comparison circuit 150, the VSET comparison circuit 267may include an amplification circuit to perform the functions asdescribed above. In one implementation, for example, the comparisoncircuit 267 may include a differential amplifier. In anotherimplementation, for example, the comparison circuit 267 may include anoperational amplifier with an output stage having a class AB bias.Regardless of the type of amplification circuit deployed by the VSETcomparison circuit 267, the VSET comparison circuit 267 is configured todetect a difference between the predetermined set voltage (V_(SETP)) andthe monitoring set voltage (V_(SETM)) and amplify the detecteddifference under a substantially linear function. The detecteddifference can be used as an indicator of the magnitude of the detectedload current 113 being below the maximum value (or peak value) of theload current 113.

Accordingly, the predetermined set voltage (V_(SETP)) can be configuredbased on a function of the set resistor 266 and an estimated maximumvalue of the output current 122, which corresponds to the maximum valueof the load current 113. The purpose of the predetermined set voltage(V_(SETP)) is to provide a comparison threshold for the monitoringcurrent path 160, such that the monitoring current path 160 can adjustthe current compensation signal 132 in response to the change of outputcurrent 122. In one implementation, for example, the predetermined setvoltage (V_(SETP)) can be configured according to Equation 1 below.V _(SETP) =I _(MON) *R _(SET)  Eq. (1)

The monitoring current (I_(MON)) 165 is a function of the supply current(I_(SUPP)) 174. The maximum value of the supply current I_(SUPP(Max))174 corresponds to the maximum value (or peak value) of the load current113 when the variable load circuit 110 has the minimum amount ofimpedance. The magnitudes of these two currents 165 and 174 arecorrelated by their respective transistor sizes. In a configurationwhere the p-channel transistor 261 has a channel width of W_(P1) and thep-channel transistor 271 has a channel width of W_(P2), thepredetermined set voltage (V_(SETP)) can be expressed by Equation 2below.

$\begin{matrix}{V_{SETP} = {\frac{W_{P\; 1}}{W_{P\; 2}}\left( {I_{{SUPP}{({M\;{ax}})}}*R_{SET}} \right)}} & {{Eq}.\mspace{14mu}(2)}\end{matrix}$

Meanwhile, the supply current 174 delivered to the first output node 121and the quiescent current (I_(N)) delivered to the n-channel transistor276 come from the current (I_(P2)) conducted by the p-channel transistor271. Thus, the predetermined set voltage (V_(SETP)) can also beexpressed by Equation 3 below.

$\begin{matrix}{V_{SETP} = {\frac{W_{P\; 1}}{W_{{P\; 2}\;}}\left( {I_{P\; 2{({{Ma}\; x})}} - I_{N}} \right)*R_{SET}}} & {{Eq}.\mspace{14mu}(3)}\end{matrix}$

The maximum current (I_(P2(Max))) conducted by the p-channel transistor271 is driven by, and thus responsive to, the maximum value (or peakvalue) of the load current 113. Thus, the p-channel transistor 261, theset resistor 266, and the comparison circuit 267 are a part of a currentmonitoring feedback loop that tracks current consumption of the variableload circuit 110. This current monitoring feedback loop also helpsadjust the amount of compensation current 133 that bypasses the variableload circuit 110 in order to maintain a relatively stable ground current114, which in turn helps sustain a relatively stable low referencevoltage (V_(REFL)) at the second output node 123.

The current compensation circuit 130 is configurable by the currentcompensation signal 132 for adjusting the compensation current 133.Thus, the current compensation circuit 130 may include one or morecurrent switch that is responsive to the adjustment indicated by thecurrent compensation signal 132. In one implementation, for example, thecurrent compensation circuit 130 may include an n-channel transistor230. The n-channel transistor 230 includes a drain node 231 that iscoupled to the first output node 121, a gate node 232 that is coupled tothe monitoring output node 162 via the control terminal 131, and asource node 233 that is coupled to the second output node 123.

The gate node 232 receives the current compensation signal 132 toestablish a gate-to-source voltage (V_(GS)) across the gate node 232 andthe source node 233. The magnitude of the V_(GS) controls the amount ofcompensation current 133 conducted from the drain node 231 to the sourcenode 233. In a configuration where the n-channel transistor 230 isdeployed as the current switch, the adjustment of the compensationcurrent 133 is directly proportional to a positive V_(GS). In analternative configuration where a p-channel transistor is deployed asthe current switch (e.g., the monitoring current path 160 includes a setcurrent source above an n-channel transistor and the inputs of the VSETcomparison circuit 267 switch polarity), the adjustment of thecompensation current 133 is directly proportional to a negative V_(GS).

To further illustrate the operation of the voltage regulation circuit120 as shown in FIG. 2, the present disclosure provides two exemplaryscenarios. In a first scenario, the impedance of the variable loadcircuit 110 reduces, thereby causing the load current 113 to increase.The increased load current 113 in turn causes the low reference voltage(V_(REFL)) to increase at the second output node 123, while the reducedload impedance causes the high reference voltage (V_(REFH)) to drop atthe first output node 121. The VREF comparison circuit 150 (e.g., theamplifier 251 and the output stage 255) senses the high referencevoltage (V_(REFH)) drop by comparing the voltage of the feedback path125 to the input reference voltage (V_(REF)).

In response, the VREF comparison circuit 150 adjust the first and secondfeedback control signal 141 and 142 to increase the supply current 174conducted by the supply current path 170. More specifically, the firstfeedback control signal 141 is reduced to increase the V_(GS) of thep-channel transistor 271, and the second feedback control signal 142 isreduced to reduce the VGS of the n-channel transistor 276. As a result,more supply current 174 is delivered as the n-channel transistor 276takes up less current. Instantaneously before the current compensationsignal 132 becomes responsive to the change of load current 113, theincreased supply current 174 allows more output current 122 to bedistributed to the variable load circuit 110 while the compensationcurrent 133 remains unchanged at that point of time. This increased loadcurrent 113 increases the high reference voltage (V_(REFH)) above thereduced impedance of the variable load circuit 110, thereby stabilizingthe voltage level at the first output node 121.

The reduction of the first feedback control signal 141 also increasesthe V_(GS) of the p-channel transistor 261 of the monitoring currentpath 160. As a result, the p-channel transistor 261 increases the amountof monitoring current 165 going through the set resistor 266, therebycausing the monitoring set voltage (V_(SETM)) to rise. The VSETcomparison circuit 267 senses the monitoring set voltage (V_(SETM))increase by comparing it against the predetermined set voltage(V_(SETP)). In response, the VSET comparison circuit 267 adjust thecurrent compensation signal 132 to reduce the amount of compensationcurrent 133 conducted by the current compensation circuit 130 so as tocompensate the increased load current 133.

In a configuration where the n-channel transistor 230 is deployed as acurrent switch, the VSET comparison circuit 267 reduces the voltage ofthe current compensation signal 132, which in turn causes the V_(GS)voltage of the n-channel transistor 230 to shrink. Accordingly, then-channel transistor 230 reduces the compensation current 133 tocompensate the increase of load current 113. In an alternativeconfiguration where a p-channel transistor is deployed as a currentswitch, the VSET comparison circuit 267 increases the voltage of thecurrent compensation signal 132 to reduce the V_(GS) voltage of thep-channel transistor. Likewise, the p-channel transistor reduces thecompensation current 133 to compensate the increase of load current 113.

In a second scenario, the impedance of the variable load circuit 110increases, thereby causing the load current 113 to decrease. Thedecrease load current 113 in turn causes the low reference voltage(V_(REFL)) to decrease at the second output node 123, while theincreased load impedance causes the high reference voltage (V_(REFH)) torise at the first output node 121. The VREF comparison circuit 150(e.g., the amplifier 251 and the output stage 255) senses the highreference voltage (V_(REFH)) rise by comparing the voltage of thefeedback path 125 to the input reference voltage (V_(REF)).

In response, the VREF comparison circuit 150 adjust the first and secondfeedback control signal 141 and 142 to reduce the supply current 174conducted by the supply current path 170. More specifically, the firstfeedback control signal 141 is increased to reduce the V_(GS) of thep-channel transistor 271, and the second feedback control signal 142 isincreased to increase the V_(GS) of the n-channel transistor 276. As aresult, less supply current 174 is delivered as the n-channel transistor276 takes up more current. Instantaneously before the currentcompensation signal 132 becomes responsive to the change of load current113, the reduced supply current 174 allows less output current 122 to bedistributed to the variable load circuit 110 while the compensationcurrent 133 remains unchanged at that point of time. This reduced loadcurrent 113 reduces the high reference voltage (V_(REFH)) above theincreased impedance of the variable load circuit 110, therebystabilizing the voltage level at the first output node 121.

The increment of the first feedback control signal 141 also reduces theVGS of the p-channel transistor 261 of the monitoring current path 160.As a result, the p-channel transistor 261 reduces the amount ofmonitoring current 165 going through the set resistor 266, therebycausing the monitoring set voltage (V_(SETM)) to drop. The VSETcomparison circuit 267 senses the monitoring set voltage (V_(SETM)) dropby comparing it against the predetermined set voltage (V_(SETP)). Inresponse, the VSET comparison circuit 267 adjust the currentcompensation signal 132 to increase the amount of compensation current133 conducted by the current compensation circuit 130 so as tocompensate the reduced load current 133.

In a configuration where the n-channel transistor 230 is deployed as acurrent switch, the VSET comparison circuit 267 increases the voltage ofthe current compensation signal 132, which in turn causes the VGSvoltage of the n-channel transistor 230 to widen. Accordingly, then-channel transistor 230 increases the compensation current 133 tocompensate the reduction of load current 113. In an alternativeconfiguration where a p-channel transistor is deployed as a currentswitch, the VSET comparison circuit 267 reduces the voltage of thecurrent compensation signal 132 to increase the VGS voltage of thep-channel transistor. Likewise, the p-channel transistor increases thecompensation current 133 to compensate the reduction of load current113.

The net effect of the compensation current 133 adjustment is to restorethe ground current 114 to its stable level such that the low referencevoltage (V_(REFL)) can remain low and stable. This feedback compensationscheme allows the total ground current 114 to return to a stabilizedlevel in a relatively short period of time. Advantageously, the voltageregulation circuit 120 as described in FIG. 2 provides a robust voltageregulation performance without introducing the typical costs of anadditional ground buffer.

FIG. 3 shows a schematic view of an exemplary voltage regulation circuit120 according to another aspect of the present disclosure. The voltageregulation circuit 120 as shown in FIG. 3 provides a specificimplementation of the voltage regulation circuit 120 as shown in FIG. 1.While the circuitry in FIG. 3 further elaborates the structure andoperations of the voltage regulation circuit 120 in FIG. 1, it does notlimit or restrict the description of FIG. 1. For instance, the circuitryin FIG. 3 provides an alternative to the circuitry in FIG. 2 while thesetwo options do not necessarily exclude each other.

In several aspects, the voltage regulation circuit 120 of FIG. 3 issimilar to the voltage regulator 120 of FIG. 2. For example, FIG. 3shows the same VREF comparison circuit 150 and the same supply currentpath 170 as FIG. 2. The voltage regulation circuit 120 of FIG. 3,however, deviates from that of FIG. 2 in two other aspects. First, themonitoring current path 160 is modified to replace the VSET comparisoncircuit 267 and the set resistor 266 with a set current source 366. Theset current source 366 is coupled with the drain node 264 of thep-channel transistor 261 via the monitoring output node 162. The setcurrent source 366 can be configured as a part of a current mirrorarrangement for conducting a set current (I_(SET)) 365. In thisparticular configuration, the monitoring current path 160 is biased toconduct a constant current, which is the set current (I_(SET)) 365regardless of the VGS voltage across the p-channel transistor 261. Tomaintain a constant current by the p-channel transistor 261, voltage ofthe drain node 264 moves in opposite direct as the voltage of the gatenode 263, which is driven by the first feedback control signal 141.

Thus, when the VREF comparison circuit 150 reduces the voltage of thefirst feedback control signal 141 in response to a voltage drop in thefirst output node 121, the voltage of the drain node 264 is pulled upsuch that the p-channel transistor 261 can maintain the monitoringcurrent 165 at a value set by the set current (I_(SET)) 365. Theincreased voltage at the drain node 264 then drives the currentcompensation signal 132 that is delivered by the monitoring output node162. Similarly, when the VREF comparison circuit 150 increases thevoltage of the first feedback control signal 141 in response to avoltage rise in the first output node 121, the voltage of the drain node264 is pulled down such that the p-channel transistor 261 can maintainthe monitoring current 165 at a value set by the set current (I_(SET))365. The decreased voltage at the drain node 264 then drives the currentcompensation signal 132 that is delivered by the monitoring output node162.

Like the predetermined set voltage (V_(SETP)), the set current (I_(SET))is predetermined based on a function of an estimated maximum value ofthe output current 122. The maximum value of the supply currentI_(SUPP(Max)) 174 corresponds to the peak of the load current 113 whenthe variable load circuit 110 is configured to incur the minimum amountof impedance. The magnitudes of these two currents 165 and 174 arecorrelated by their respective transistor size. In a configuration wherethe p-channel transistor 261 has a channel width of W_(P1) and thep-channel transistor 271 has a channel width of W_(P2), thepredetermined set current (I_(SET)) can be expressed by Equation 4below.

$\begin{matrix}{I_{SET} = {\frac{W_{P\; 1}}{W_{P\; 2}}\left( I_{{SUPP}{({{Ma}\; x})}} \right)}} & {{Eq}.\mspace{14mu}(4)}\end{matrix}$

Because the supply current 174 and the quiescent current (I_(N))conducted by the n-channel transistor 276 come from the current (I_(P2))conducted by the p-channel transistor 271, the predetermined set current(I_(SET)) can also be expressed by Equation 5 below.

$\begin{matrix}{I_{SET} = {\frac{W_{P\; 1}}{W_{{P\; 2}\;}}\left( {I_{P\; 2{({M\;{ax}})}} - I_{N}} \right)}} & {{Eq}.\mspace{14mu}(5)}\end{matrix}$

The maximum current (I_(P2(Max))) conducted by the p-channel transistor271 is driven by, and thus responsive to, the peak of the load current113. Thus, the p-channel transistor 261 and the set current source 366are a part of a current monitoring feedback loop that tracks currentconsumption of the variable load circuit 110. This current monitoringfeedback loop also helps adjust the amount of compensation current 133that bypasses the variable load circuit 110 in order to maintain arelatively stable ground current 114, which in turn helps sustain arelatively stable low reference voltage (VREFL) at the second outputnode 123.

The second modification of FIG. 3 involves the current compensationcircuit 130. Similar to FIG. 2, the current compensation circuit 130 inFIG. 3 is configurable by the current compensation signal 132 forconducting the compensation current 133. Thus, the current compensationcircuit 130 may include one or more current switch that is responsive tothe adjustment indicated by the current compensation signal 132. Butbecause the polarity of the current compensation signal 132 has flippedwhen compared to the current compensation signal 132 as described inFIG. 2, the current compensation circuit 130 may deploy one or morep-channel transistor 330 to implement the feedback compensation schemeas described in FIG. 2. The p-channel transistor 330 includes a sourcenode 331 that is coupled to the first output node 121, a gate node 332that is coupled to the monitoring output node 162 via the controlterminal 131, and a drain node 333 that is coupled to the second outputnode 123.

The gate node 332 receives the current compensation signal 132 toestablish a gate-to-source voltage (V_(GS)) across the gate node 332 andthe source node 331. The magnitude of the V_(GS) controls the amount ofcompensation current 133 conducted from the source node 331 to the drainnode 333. In a configuration where the p-channel transistor 330 isdeployed as the current switch, the adjustment of the compensationcurrent 133 is directly proportional to a negative VGS. In analternative configuration where an n-channel transistor is deployed asthe current switch (e.g., the monitoring current path 160 includes a setcurrent source above an n-channel transistor), the adjustment of thecompensation current 133 is directly proportional to a positive VGS.

To further illustrate the operation of the voltage regulation circuit120 as shown in FIG. 3, the present disclosure provides two exemplaryscenarios. In a first scenario, the impedance of the variable loadcircuit 110 reduces, thereby causing the load current 113 to increase.The increased load current 113 in turn causes the low reference voltage(V_(REFL)) to increase at the second output node 123, while the reducedload impedance causes the high reference voltage (V_(REFH)) to drop atthe first output node 121. The VREF comparison circuit 150 (e.g., theamplifier 251 and the output stage 255) senses the high referencevoltage (V_(REFH)) drop by comparing the voltage of the feedback path125 to the input reference voltage (V_(REF)).

In response, the VREF comparison circuit 150 adjust the first and secondfeedback control signal 141 and 142 to increase the supply current 174conducted by the supply current path 170. More specifically, the firstfeedback control signal 141 is reduced to increase the VGS of thep-channel transistor 271, and the second feedback control signal 142 isreduced to reduce the VGS of the n-channel transistor 276. As a result,more supply current 174 is delivered as the n-channel transistor 276takes up less current. Instantaneously before the current compensationsignal 132 becomes responsive to the change of load current 113, theincreased supply current 174 allows more output current 122 to bedistributed to the variable load circuit 110 while the compensationcurrent 133 remains unchanged at that point of time. This increased loadcurrent 113 increases the high reference voltage (V_(REFH)) above thereduced impedance of the variable load circuit 110, thereby stabilizingthe voltage level at the first output node 121.

The reduction of the first feedback control signal 141 also increasesthe VGS of the p-channel transistor 261 of the monitoring current path160. As a result, the p-channel transistor 261 increases the voltage ofthe drain node 264, thereby causing the voltage of the currentcompensation signal 132 to rise. In response, the current compensationcircuit 130 reduces the amount of compensation current 133 conducted bythe current compensation circuit 130 to compensate the increased loadcurrent 133.

For example, in a configuration where the p-channel transistor 330 isdeployed as a current switch, the increased voltage of the currentcompensation signal 132 causes the VGS voltage of the p-channeltransistor 330 to shrink. Accordingly, the p-channel transistor 330reduces the compensation current 133 to compensate the increase of loadcurrent 113. In an alternative configuration where an n-channeltransistor is deployed as a current switch, the voltage of the currentcompensation signal 132 is decreased to shrink the VGS voltage of then-channel transistor. Likewise, the n-channel transistor reduces thecompensation current 133 to compensate the increase of load current 113.

In a second scenario, the impedance of the variable load circuit 110increases, thereby causing the load current 113 to decrease. Thedecrease load current 113 in turn causes the low reference voltage(V_(REFL)) to decrease at the second output node 123, while theincreased load impedance causes the high reference voltage (V_(REFH)) torise at the first output node 121. The VREF comparison circuit 150(e.g., the amplifier 251 and the output stage 255) senses the highreference voltage (V_(REFH)) rise by comparing the voltage of thefeedback path 125 to the input reference voltage (V_(REF)).

In response, the VREF comparison circuit 150 adjust the first and secondfeedback control signal 141 and 142 to reduce the supply current 174conducted by the supply current path 170. More specifically, the firstfeedback control signal 141 is increased to reduce the V_(GS) of thep-channel transistor 271, and the second feedback control signal 142 isincreased to increase the V_(GS) of the n-channel transistor 276. As aresult, less supply current 174 is delivered as the n-channel transistor276 takes up more current. Instantaneously before the currentcompensation signal 132 becomes responsive to the change of load current113, the reduced supply current 174 allows less output current 122 to bedistributed to the variable load circuit 110 while the compensationcurrent 133 remains unchanged at that point of time. This reduced loadcurrent 113 reduces the high reference voltage (V_(REFH)) above theincreased impedance of the variable load circuit 110, therebystabilizing the voltage level at the first output node 121.

The increment of the first feedback control signal 141 also reduces theVGS of the p-channel transistor 261 of the monitoring current path 160.As a result, the p-channel transistor 261 reduces the voltage of thedrain node 264, thereby causing the voltage of the current compensationsignal 132 to drop. In response, the current compensation circuit 130increases the amount of compensation current 133 conducted by thecurrent compensation circuit 130 to compensate the decreased loadcurrent 133.

For example, in a configuration where the p-channel transistor 330 isdeployed as a current switch, the decreased voltage of the currentcompensation signal 132 causes the V_(GS) voltage of the p-channeltransistor 330 to widen. Accordingly, the p-channel transistor 330increases the compensation current 133 to compensate the reduction ofload current 113. In an alternative configuration where an n-channeltransistor is deployed as a current switch, the voltage of the currentcompensation signal 132 is increase to widen the V_(GS) voltage of then-channel transistor. Likewise, the n-channel transistor increases thecompensation current 133 to compensate the reduction of load current113.

Like the operation as described in FIG. 2, the net effect of thecompensation current 133 adjustment is to restore the ground current 114to its stable level such that the low reference voltage (V_(REFL)) canremain low and stable. This feedback compensation scheme allows thetotal ground current 114 to return to a stabilized level in a relativelyshort period of time. Advantageously, the voltage regulation circuit 120as described in FIG. 3 provides a robust voltage regulation performancewithout introducing the typical costs of an additional ground buffer.Moreover, for not including the VSET comparison circuit 267, the voltageregulation circuit 120 of FIG. 3 consumes less power and takes up lessspace than the voltage regulation circuit 120 of FIG. 2.

FIG. 4 shows a schematic view of an exemplary voltage regulation circuit120 according to yet another aspect of the present disclosure. Thevoltage regulation circuit 120 as shown in FIG. 4 provides a specificimplementation of the voltage regulation circuit 120 as shown in FIG. 1.While the circuitry in FIG. 4 further elaborates the structure andoperations of the voltage regulation circuit 120 in FIG. 1, it does notlimit or restrict the description of FIG. 1. For instance, the circuitryin FIG. 4 provides an alternative to the circuitries in FIG. 2 and FIG.3 while these options do not necessarily exclude one another.

In several aspects, the voltage regulation circuit 120 of FIG. 4 issimilar to the voltage regulator 120 of FIG. 1. For example, FIG. 4shows a current sensor 140 operating in conjunction with a currentcompensation circuit 130 according to the general description of FIG. 1.More specifically, the voltage regulation circuit 120 of FIG. 4 providesthe same feedback compensation mechanism to sustain a relatively stableground current 114 across the ground parasitic resistor 115. The voltageregulation circuit 120 of FIG. 4, however, deviates from theimplementations of FIGS. 2 and 3 in two aspects. First the voltageregulation circuit 120 of FIG. 4 deploys a two-stage sensing approachinstead of a duo-current-path approach. Second, the voltage regulationcircuit 120 of FIG. 4 incorporates the VREF comparison circuit 150 intoone of the sensing stage to simplify the current sensing and adjustmentcircuitry.

In this particular configuration, the current sensor 140 includes afirst sensing stage 410 and a second sensing stage 430. The firstsensing stage 410 is coupled with the first output node 121 for sensinga change of the output current 122. Based on this feedback sensingmechanism, the first sensing stage 410 generates a current sense signalthat responds positively to the sensed change of the output current 122.The first sensing stage 410 includes a VREF comparison circuit 411 and areference resistor 416. The VREF comparison circuit 411 includes anegative input lead 412 that is coupled to the first output node 121 anda positive input lead 413 that is coupled to the input node 124.

Similar to the VREF comparison circuit 150, the VREF comparison circuit411 serves as a tracking means for tracking a change of voltage at thefirst output node 121. In general, the VREF comparison circuit 411 isconfigured to track the high reference voltage (V_(REFH)) at the firstoutput node 121 by comparing the high reference voltage (V_(REFH)) tothe input reference voltage (V_(REF)) received by the input node 124 ofthe voltage regulation circuit 120. Based on this comparison, the VREFcomparison circuit 411 generates current sense signal at its output lead414. The current sense signal has a voltage level that is high then thehigh reference voltage (V_(REFH)) at the first output node 121. This isbecause the current sense signal serves to provide the supply current174 that flows through the reference resistor 416. The supply current174 establishes a potential drop (i.e., V_(REFD)) across an output lead414 of the VREF comparison circuit 411 and the first output node 121 asthe reference resistor 416 is coupled between these two nodes. When theVREF comparison circuit 411 operates within its operation range, thecurrent sense signal may have a substantially linear relationship with adifference between the input reference voltage (V_(REF)), which isreceived from the input node 124, and the high reference voltage(V_(REFL)), which is received from the first output node 121.

Like the VREF comparison circuit 150, the VREF comparison circuit 411may include an amplifier that detects a difference between the inputreference voltage (V_(REF)) and the high reference voltage (V_(REFH))and amplifies the detected difference under a substantially linearfunction, for instance: V_(CS)=Ao*(V_(plus)−V_(minus)), where V_(CS) isthe voltage level of the current sense signal at the output lead 414 andAo is the gain of the linear function. The amplifier of the VREFcomparison circuit 411 may include one or more differential amplifier inone implementation. In another implementation, the amplifier of the VREFcomparison circuit 411 may include one or more operational amplifier. Inyet another implementation, the amplifier of the VREF comparison circuit411 can include other types of amplification circuits that are suitablefor performing the functions as described above. To keep the voltage atthe first output node 121 at the higher reference voltage (V_(REFH)),the voltage level V_(CS) of the current sense signal is regulated by afunction of the supply current (I_(SUPP)) 174 and the reference resistor(R_(REF)) 416. For example, the voltage level V_(CS) of the currentsense signal can be expressed by Equation 6 below.V _(CS) =I _(SUPP) *R _(REF) +V _(REF)  Eq. (6)

The second sensing stage 430 is coupled to the first sensing stage 410to receive the current sense signal. The second sensing stage 430 servesa monitoring function that is similar to that of the monitoring currentpath 160. For instance, the second sensing stage 430 includes a VSETcomparison circuit 431 that is similar to the VSET comparison circuit267. The VSET comparison circuit 431 is configured to generate a currentcompensation signal 441 that is based on a comparison between thecurrent sense signal and a predetermined set voltage (V_(SETP)).

The VSET comparison circuit 431 serves as a tracking means for trackinga change in the supply current 174, which is proportional to a change inthe output current 122. The VSET comparison circuit 431 includes a firstinput 432 (which can be a negative input), a second input 435 (which canbe a positive input), and a comparison output 434. The first input 432is coupled to the output lead 414 to receive the current sense voltage(V_(CS)) of the current sense signal, whereas the second input 435 iscoupled to a voltage source to receive the predetermined set voltage(V_(SETP)). The VSET comparison circuit 431 delivers the currentcompensation signal 441 to the current compensation circuit 130 via itsoutput lead 434. When the VSET comparison circuit 431 operates withinits operation range, the current compensation signal 441 may have asubstantially linear relationship with a difference between thepredetermined set voltage (V_(SETP)) and the current sense voltage(V_(CS)).

The VSET comparison circuit 431 may include an amplification circuit toperform the functions as described above. In one implementation, forexample, the comparison circuit 431 may include a differentialamplifier. In another implementation, for example, the comparisoncircuit 431 may include an operational amplifier with an output stagehaving a class AB bias. Regardless of the type of amplification circuitdeployed by the VSET comparison circuit 431, the VSET comparison circuit431 is configured to detect a difference between the predetermined setvoltage (V_(SETP)) and the current sense voltage (V_(CS)) and amplifythe detected difference under a substantially linear function.

The predetermined set voltage (V_(SETP)) is configured based on afunction of the maximum current sense voltage (V_(CSMax)). The purposeof the predetermined set voltage (VSETP) is to provide a comparisonthreshold for the supply current 174, such that the second sensing stage430 can adjust the current compensation signal 441 in response to thechange of output current 122. In one implementation, the predeterminedset voltage (V_(SETP)) can be configured according to Equation 7 below.V _(SETP) =I _(SUPPMax) *R _(REF) +V _(REF)  Eq. (7)

The supply current (I_(SUPP)) 174 can be reserved to accommodate themaximum (or peak) draw of the output current 122 which is triggered by asurge of the load current 113 when the impedance of the variable loadcircuit 110 reaches its minimum value. Thus, this maximum supply current(I_(SUPPMax)) forms the baseline for comparing the fluctuating supplycurrent (I_(SUPP)) 174. Because of the headroom established by theI_(SUPPMax)*R_(REF) term, the predetermined set voltage (V_(SET)) isconfigured to be greater than the input reference voltage (V_(REF)). Andto reduce the impact of this headroom, the reference resistor 416 mayhave a resistance that limits the headroom to about 100 mV.

To further illustrate the operation of the voltage regulation circuit120 as shown in FIG. 4, the present disclosure provides two exemplaryscenarios. In a first scenario, the impedance of the variable loadcircuit 110 reduces, thereby causing the load current 113 to increase.The increased load current 113 in turn causes the low reference voltage(V_(REFL)) to increase at the second output node 123, while the reducedload impedance causes the high reference voltage (V_(REFH)) to drop atthe first output node 121. The VREF comparison circuit 411 senses thehigh reference voltage (V_(REFH)) drop by comparing the voltage fed backfrom the first output node 121 to the input reference voltage (V_(REF)).

In response, the VREF comparison circuit 411 adjust the current sensesignal to increase the supply current 174 conducted by the referenceresistor 416 by raising the current sense voltage V_(CS) at its outputlead 414. As a result, more supply current 174 is delivered before thecurrent compensation signal 441 becomes responsive to the change of loadcurrent 113. The increased supply current 174 allows more output current122 to be distributed to the variable load circuit 110 while thecompensation current 133 remains unchanged at that point of time. Thisincreased load current 113 increases the high reference voltage(V_(REFH)) to approach the reference input voltage (V_(REF)), therebystabilizing the voltage level at the first output node 121. The VSETcomparison circuit 431 senses the current sense voltage (V_(CS))increase by comparing it against the predetermined set voltage(V_(SETP)). In response, the VSET comparison circuit 431 adjust thecurrent compensation signal 441 to reduce the amount of compensationcurrent 133 conducted by the current compensation circuit 130.

In a configuration where the n-channel transistor 230 is deployed as acurrent switch, the VSET comparison circuit 431 reduces the voltage ofthe current compensation signal 132, which in turn causes the VGSvoltage of the n-channel transistor 230 to shrink. Accordingly, then-channel transistor 230 reduces the compensation current 133 tocompensate the increase of load current 113. In an alternativeconfiguration where a p-channel transistor is deployed as a currentswitch, the VSET comparison circuit 431 increases the voltage of thecurrent compensation signal 132 to reduce the VGS voltage of thep-channel transistor. Likewise, the p-channel transistor reduces thecompensation current 133 to compensate the increase of load current 113.

In a second scenario, the impedance of the variable load circuit 110increases, thereby causing the load current 113 to decrease. Thedecreased load current 113 in turn causes the low reference voltage(V_(REFL)) to decrease at the second output node 123, while theincreased load impedance causes the high reference voltage (V_(REFH)) torise at the first output node 121. The VREF comparison circuit 411senses the high reference voltage (V_(REFH)) rise by comparing thevoltage fed back from the first output node 121 to the input referencevoltage (V_(REF)).

In response, the VREF comparison circuit 411 adjust the current sensesignal to reduce the supply current 174 conducted by the referenceresistor 416 by dropping the current sense voltage V_(CS) at its outputlead 414. As a result, less supply current 174 is delivered before thecurrent compensation signal 441 becomes responsive to the change of loadcurrent 113. The reduced supply current 174 allows less output current122 to be distributed to the variable load circuit 110 while thecompensation current 133 remains unchanged at that point of time. Thisincreased load current 113 reduces the high reference voltage (V_(REFH))to approach the reference input voltage (V_(REF)), thereby stabilizingthe voltage level at the first output node 121. The VSET comparisoncircuit 431 senses the current sense voltage (V_(CS)) reduction bycomparing it against the predetermined set voltage (V_(SETP)). Inresponse, the VSET comparison circuit 431 adjust the currentcompensation signal 441 to increase the amount of compensation current133 conducted by the current compensation circuit 130.

In a configuration where the n-channel transistor 230 is deployed as acurrent switch, the VSET comparison circuit 431 increases the voltage ofthe current compensation signal 132, which in turn causes the V_(GS)voltage of the n-channel transistor 230 to widen. Accordingly, then-channel transistor 230 increases the compensation current 133 tocompensate the reduction of load current 113. In an alternativeconfiguration where a p-channel transistor is deployed as a currentswitch, the VSET comparison circuit 431 reduces the voltage of thecurrent compensation signal 132 to increase the V_(GS) voltage of thep-channel transistor. Likewise, the p-channel transistor increases thecompensation current 133 to compensate the reduction of load current113.

Like the operation as described in FIG. 2, the net effect of thecompensation current 133 adjustment is to restore the ground current 114to its stable level such that the low reference voltage (V_(REFL)) canremain low and stable. This feedback compensation scheme allows thetotal ground current 114 to return to a stabilized level in a relativelyshort period of time. Advantageously, the voltage regulation circuit 120as described in FIG. 4 provides a robust voltage regulation performancewithout introducing the typical costs of an additional ground buffer.Moreover, for not including the monitoring current path 160 and thesupply current path 170, the voltage regulation circuit 120 of FIG. 4consumes less power than the voltage regulation circuit 120 of FIGS. 2and 3.

FIG. 5 shows a schematic view of an exemplary digital-to-analogconversion (DAC) circuit 500 according to an aspect of the presentdisclosure. The DAC circuit 500 can be included as a part of thevariable load circuit 110 as shown in FIG. 1 and as discussed along withFIGS. 2-4. When any one of the voltage regulation circuits 120 (see,e.g., FIGS. 1-4) is used in conjunction with the DAC circuit 500, thecombined circuitry forms a digital-to-analog conversion system, whichcan be implemented in a single integrated circuit or as discretecomponents disposed on a printed circuit board. At its periphery, theDAC circuit 500 includes a high reference input port 501, a lowreference input port 502, a digital code input port 503, and an analogoutput port 504.

The high reference input port 501 is coupled to the high reference node111 to receive the high reference voltage (V_(REFH)), whereas the lowreference input port 502 is coupled to the low reference node 112 toreceive the low reference voltage (V_(REFL)). The digital code inputport 503 is configured to receive a digital code from which the DACcircuit 500 converts to an analog output for delivery at its analogoutput port 504. This conversion is performed by the internalcircuitries of the DAC circuit 500, including a variable resistancenetwork 510, an amplifier 540, and a feedback resistor 542. The variableresistance network 510 may exhibit variable resistance based on thedigital code received at the digital code input port 503.

The variable resistance network 510 includes a resistor ladder (e.g., anR2R ladder) 520 and a DAC switch circuit 530. The DAC switch circuit 530is controlled by the digital code such that it may selectively connectone or more branches (e.g., the 2R branch) of the resistor ladder 520 toeither the high reference input port 501 or the low reference input port502. This selective connection may result in a change of impedanceacross the high reference input port 501 and the low reference inputport 502. To that end, the load current 113 delivered across these twoports (i.e., 501 and 502) is also dependent on the digital code receivedby the digital code input port 504.

A few embodiments have been described in detail above, and variousmodifications are possible. The disclosed subject matter, including thefunctional operations described in this specification, can beimplemented in electronic circuitry, computer hardware, firmware,software, or in combinations of them, such as the structural meansdisclosed in this specification and structural equivalents thereof,including potentially a program operable to cause one or more dataprocessing apparatus to perform the methods and/or operations described(such as a program encoded in a computer-readable medium, which can be amemory device, a storage device, a machine-readable storage substrate,or other physical, machine-readable medium, or a combination of one ormore of them).

Consistent with the present disclosure, the term “configured to”purports to describe the structural and functional characteristics ofone or more tangible non-transitory components. For example, the term“configured to” can be understood as having a particular configurationthat is designed or dedicated for performing a certain function. Withinthis understanding, a device is “configured to” perform a certainfunction if such a device includes tangible non-transitory componentsthat can be enabled, activated, or powered to perform that certainfunction. While the term “configured to” may encompass the notion ofbeing configurable, this term should not be limited to such a narrowdefinition. Thus, when used for describing a device, the term“configured to” does not require the described device to be configurableat any given point of time.

While this specification contains many specifics, these should not beconstrued as limitations on the scope of what may be claimed, but ratheras descriptions of features that may be specific to particularembodiments. Certain features that are described in this specificationin the context of separate embodiments can also be implemented incombination in a single embodiment. Conversely, various features thatare described in the context of a single embodiment can also beimplemented in multiple embodiments separately or in any suitablesubcombination. Moreover, although features may be described above asacting in certain combinations and even initially claimed as such, oneor more features from a claimed combination can in some cases be excisedfrom the combination, and the claimed combination may be directed to asubcombination or variation of a subcombination.

Similarly, while operations are depicted in the drawings in a particularorder, this should not be understood as requiring that such operationsbe performed in the particular order shown or in sequential order, orthat all illustrated operations be performed, to achieve desirableresults unless such order is recited in one or more claims. In certaincircumstances, multitasking and parallel processing may be advantageous.Moreover, the separation of various system components in the embodimentsdescribed above should not be understood as requiring such separation inall embodiments.

What is claimed is:
 1. A voltage regulation device, comprising: firstand second output nodes; a current compensation circuit including: acurrent input node coupled to the first output node; a current outputnode coupled to the second output node; and a control terminal; and acurrent sensor circuit including: a supply current path having a supplyoutput node coupled to the first output node and the current input node;and a monitoring current path having a monitoring output node coupled tothe control terminal.
 2. The voltage regulation device of claim 1,wherein: the supply current path includes a first p-channel transistorarranged along a feedback path originating from the first output nodeand positioned upstream of the current output node; and the monitoringcurrent path includes a second p-channel transistor arranged along thefeedback path originating from the first output node and positionedupstream of the monitoring output node.
 3. The voltage regulation deviceof claim 2, wherein the monitoring current path includes: a resistorcoupled with the second p-channel transistor at a set node; and acomparison circuit having: a first input coupled to the set node; asecond input; and a comparison output coupled to the monitoring outputnode.
 4. The voltage regulation device of claim 3, wherein the currentcompensation circuit includes an n-channel transistor having: a drainnode coupled to the current input node; a gate node coupled to thecontrol terminal; and a source node coupled to the current output node.5. The voltage regulation device of claim 2, wherein the monitoringcurrent path includes: a current source coupled to the second p-channeltransistor at the monitoring output node.
 6. The voltage regulationdevice of claim 5, wherein the current compensation circuit includes anp-channel transistor having: a source node coupled to the current inputnode; a gate node coupled to the control terminal; and a drain nodecoupled to the current output node.
 7. A voltage regulation circuit,comprising: first and second output nodes; a current compensationcircuit including: a current input node coupled to the first outputnode; a current output node coupled to the second output node; and acontrol terminal; and a current sensor circuit including: a firstsensing stage establishing a feedback path with the first output node;and a second sensing stage having an input coupled to the first sensingstage, and an output coupled to the control terminal of the currentcompensation circuit.
 8. The voltage regulation circuit of claim 7,wherein the first sensing stage includes: a comparison circuit having anoutput lead, a positive input lead, and a negative input lead coupledwith the first output node to establish the feedback path; and aresistor coupled in series between the output lead and the first outputnode.
 9. The voltage regulation circuit of claim 7, wherein the secondsensing stage includes: a comparison circuit having a negative inputlead of the input coupled to the first sensing stage, a positive inputlead of the input, and an output lead of the output coupled to thecontrol terminal of the current compensation circuit.
 10. The voltageregulation circuit of claim 7, wherein the current compensation circuitincludes an n-channel transistor having: a drain node coupled to thecurrent input node; a gate node coupled to the control terminal; and asource node coupled to the current output node.
 11. A digital-to-analogconversion (DAC) system, comprising: a variable resistance networkhaving first and second reference nodes, and a load positionedtherebetween and adjustable based on a digital code; a currentcompensation circuit including: a current input node coupled to thefirst output node; a current output node coupled to the second outputnode; and a control terminal; and a current sensor circuit establishinga feedback path with the first reference node, and having an outputcoupled to the control terminal.
 12. The DAC system of claim 11, whereinthe current sensor circuit includes: a supply current path having asupply output node coupled to the first reference node and the currentinput node; and a monitoring current path having a monitoring outputnode coupled to the control terminal of the current compensationcircuit.
 13. The DAC system of claim 12, wherein: the supply currentpath includes a first p-channel transistor arranged along the feedbackpath originating from the first reference node and positioned upstreamof the current output node; and the monitoring current path includes asecond p-channel transistor arranged along the feedback path originatingfrom the first reference node and positioned upstream of the monitoringoutput node.
 14. The DAC system of claim 13, wherein the monitoringcurrent path includes: a resistor coupled with the second p-channeltransistor at a set node; and a comparison circuit having: a first inputcoupled to the set node; a second input; and a comparison output coupledto the monitoring output node.
 15. The DAC system of claim 14, whereinthe current compensation circuit includes an n-channel transistorhaving: a drain node coupled to the current input node; a gate nodecoupled to the control terminal; and a source node coupled to thecurrent output node.
 16. The DAC system of claim 13, wherein themonitoring current path includes: a current source coupled to the secondp-channel transistor at the monitoring output node.
 17. The DAC systemof claim 16, wherein the current compensation circuit includes anp-channel transistor having: a source node coupled to the current inputnode; a gate node coupled to the control terminal; and a drain nodecoupled to the current output node.
 18. The DAC system of claim 11,wherein the current sensor circuit includes: a first sensing stageestablishing the feedback path with the first reference node; and asecond sensing stage having an input coupled to the first sensing stage,and the output coupled to the control terminal of the currentcompensation circuit.
 19. The DAC system of claim 18, wherein the firstsensing stage includes: a comparison circuit having an output lead, apositive input lead, and a negative input lead coupled with the firstreference node to establish the feedback path; and a resistor coupled inseries between the output lead and the first reference node.
 20. The DACsystem of claim 18, wherein the second sensing stage includes: acomparison circuit having a negative input lead of the input coupled tothe first sensing stage, a positive input lead of the input, and anoutput lead of the output coupled to the control terminal of the currentcompensation circuit.